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FSDH321, FSDL321
Features
* Internal Avalanche Rugged Sense FET * Consumes only 0.65W at 240VAC & 0.3W load with Advanced Burst-Mode Operation * Frequency Modulation for low EMI * Precision Fixed Operating Frequency * Internal Start-up Circuit * Pulse by Pulse Current Limiting * Abnormal Over Current Protection * Over Voltage Protection * Over Load Protection * Internal Thermal Shutdown Function * Auto-Restart Mode * Under Voltage Lockout * Low Operating Current (max 3mA) * Adjustable Peak Current Limit * Built-in Soft Start
Green Mode Fairchild Power Switch (FPSTM)
OUTPUT POWER TABLE
230VAC 15%(3) PRODUCT FSDL321 FSDH321 FSDL0165RN FSDM0265RN FSDH0265RN FSDL0365RN FSDM0365RN FSDL321L FSDH321L FSDL0165RL FSDM0265RL FSDH0265RL FSDL0365RL FSDM0365RL Adapter(1) 11W 11W 13W 16W 16W 19W 19W 11W 11W 13W 16W 16W 19W 19W Open Frame(2) 17W 17W 23W 27W 27W 30W 30W 17W 17W 23W 27W 27W 30W 30W 85-265VAC Adapter(1) 8W 8W 11W 13W 13W 16W 16W 8W 8W 11W 13W 13W 16W 16W Open Frame(2) 12W 12W 17W 20W 20W 24W 24W 12W 12W 17W 20W 20W 24W 24W
Applications
* SMPS for STB, Low cost DVD * Auxiliary Power for PC * Adaptor for Charger
Description
The FSDx321(x stands for H, L) are integrated Pulse Width Modulators (PWM) and Sense FETs specifically designed for high performance offline Switch Mode Power Supplies (SMPS) with minimal external components. Both devices are integrated high voltage power switching regulators which combine an avalanche rugged Sense FET with a current mode PWM control block. The integrated PWM controller features include: a fixed oscillator with frequency modulation for reduced EMI, Under Voltage Lock Out (UVLO) protection, Leading Edge Blanking (LEB), optimized gate turn-on/turn-off driver, Thermal Shut Down (TSD) protection, Abnormal Over Current Protection (AOCP) and temperature compensated precision current sources for loop compensation and fault protection circuitry. When compared to a discrete MOSFET and controller or RCC switching converter solution, the FSDx321 reduce total component count, design size, weight and at the same time increase efficiency, productivity, and system reliability. Both devices are a basic platform well suited for cost effective designs of flyback converters.
Table 1. Notes: 1. Typical continuous power in a non-ventilated enclosed adapter measured at 50C ambient. 2. Maximum practical continuous power in an open frame design at 50C ambient. 3. 230 VAC or 100/115 VAC with doubler.
Typical Circuit
AC IN
DC OUT
Vstr Ipk PWM Vfb
Drain
Vcc
Source
Figure 1. Typical Flyback Application
Rev.1.0.2
(c)2004 Fairchild Semiconductor Corporation
FSDH321, FSDL321
Internal Block Diagram
Vcc 2
+
Vstr 5
Drain 6,7,8
Istart
Soft start
VBURL/VBURH
-
8V/12V VBURH
Vcc I B_PEAK
Vcc I delay Vcc
Vcc good
Freq. Modulation
Vref
Internal Bias
OSC
VFB 3
I FB
Normal
S
Q
PWM
Burst
R Q
2.5R Ipk 4 R
Gate driver LEB
V SD Vcc
S Q
1 GND AOCP Vocp
Vovp TSD Vcc good
R Q
Figure 2. Functional Block Diagram of FSDx321
2
FSDH321, FSDL321
Pin Definitions
Pin Number 1 Pin Name GND Pin Function Description Sense FET source terminal on primary side and internal control ground. Positive supply voltage input. Although connected to an auxiliary transformer winding, current is supplied from pin 5 (Vstr) via an internal switch during startup (see Internal Block Diagram section). It is not until Vcc reaches the UVLO upper threshold (12V) that the internal start-up switch opens and device power is supplied via the auxiliary transformer winding. The feedback voltage pin is the non-inverting input to the PWM comparator. It has a 0.9mA current source connected internally while a capacitor and optocoupler are typically connected externally. A feedback voltage of 6V triggers over load protection (OLP). There is a time delay while charging between 3V and 6V using an internal 5uA current source, which prevents false triggering under transient conditions but still allows the protection mechanism to operate under true overload conditions. Pin to adjust the current limit of the Sense FET. The feedback 0.9mA current source is diverted to the parallel combination of an internal 2.8k resistor and any external resistor to GND on this pin to determine the current limit. If this pin is tied to Vcc or left floating, the typical current limit will be 0.7A. This pin connects directly to the rectified AC line voltage source. At start up the internal switch supplies internal bias and charges an external storage capacitor placed between the Vcc pin and ground. Once the Vcc reaches 12V, the internal switch is disabled. The Drain pin is designed to connect directly to the primary lead of the transformer and is capable of switching a maximum of 650V. Minimizing the length of the trace connecting this pin to the transformer will decrease leakage inductance.
2
Vcc
3
Vfb
4
Ipk
5
Vstr
6, 7, 8
Drain
Pin Configuration
8DIP 8LSOP GND 1 Vcc 2 Vfb 3 Ipk 4 8 Drain 7 Drain 6 Drain 5 Vstr
Figure 3. Pin Configuration (Top View)
3
FSDH321, FSDL321
Absolute Maximum Ratings
(Ta=25C, unless otherwise specified) Parameter Maximum Vstr Pin Voltage Maximum Drain Pin Voltage Drain-Gate Voltage (RGS=1M) Gate-Source (GND) Voltage Drain Current Pulsed (1) Continuous Drain Current (Tc=25C) Continuous Drain Current (Tc=100C) Single Pulsed Avalanche Input Voltage Range Total Power Dissipation Operating Junction Temperature. Operating Ambient Temperature. Storage Temperature Range. Energy (2) Maximum Supply Voltage Symbol VSTR,MAX VDRAIN,MAX VDGR VGS IDM ID ID EAS VCC,MAX VFB PD TJ TA TSTG Value 650 650 650 20 1.5 0.7 0.32 10 20 -0.3 to Vstop 1.25 +150 -25 to +85 -55 to +150 Unit V V V V ADC ADC ADC mJ V V W C C C
Note: 1. Repetitive rating: Pulse width limited by maximum junction temperature 2. L = 24mH, starting Tj = 25C
4
FSDH321, FSDL321
Electrical Characteristics (Sense FET Part)
(Ta = 25C unless otherwise specified) Parameter Sense FET SECTION Drain-Source Breakdown Voltage Startup Voltage (Vstr) Breakdown BVDSS BVSTR VGS=0V, ID=50A VCC=0V, ID=1mA VDS=Max. Rating, VGS=0V VDS=0.8Max. Rating, VGS=0V, TC=125C VGS=10V, ID=0.5A VDS=50V, ID=0.5A VGS=0V, VDS=25V, f=1MHz VDD=0.5B VDSS, ID=1.0A (MOSFET switching time is essentially independent of operating temperature) VGS=10V, ID=1.0A, VDS=0.5B VDSS (MOSFET switching time is essentially independent of operating temperature) 650 650 1.0 720 720 14 1.3 162 18 3.8 9.5 19 33 42 7.0 3.1 0.4 25 200 19 nC ns pF V V A A S Symbol Condition Min. Typ. Max. Unit
Zero Gate Voltage Drain Current
IDSS
Static Drain-Source on Resistance
(Note)
RDS(ON) gfs CISS COSS CRSS td(on) tr td(off) tf Qg Qgs Qgd
Forward Trans conductance (Note) Input Capacitance Output Capacitance Reverse Transfer Capacitance Turn on Delay Time Rise Time Turn Off Delay Time Fall Time Total Gate Charge (Gate-Source + Gate-Drain) Gate-Source Charge Gate-Drain (Miller) Charge
Note: 1. Pulse test: Pulse width 300S, duty 2% 2. 1 S = --R
5
FSDH321, FSDL321
Electrical Characteristics (Control Part) (Continued)
(Ta=25C unless otherwise specified) Parameter UVLO SECTION Start Threshold Voltage Stop Threshold Voltage OSCILLATOR SECTION Initial Accuracy Frequency Modulation Initial Accuracy Frequency Modulation Frequency Change With Temperature (2) Maximum Duty Cycle FEEDBACK SECTION Feedback Source Current Shutdown Feedback Voltage Shutdown Delay Current BURST MODE SECTION VBURH Burst Mode Voltage VBURL Hysteresis CURRENT LIMIT(SELF-PROTECTION)SECTION Peak Current Limit(3) Current Limit Delay(1) SOFT START SECTION Soft Start Time PROTECTION SECTION Thermal Shutdown Temperature (1) Over Voltage Protection TOTAL STANDBY CURRENT SECTION Startup Charging Current Operating Supply Current (Control Part Only) ICH IOP VCC=0V VCC = 14V, Vfb = 0V 0.7 1 0.85 3 1.0 5 mA mA TSD VOVP 125 18 145 19 20 C V TSS Vfb = 4V 10 15 20 ms ILIM TCLD Tj = 25C Tj = 25C 0.60 0.70 600 0.80 A ns Tj = 25C 0.4 0.25 0.5 0.35 150 0.6 0.45 V V mV IFB VSD IDELAY Ta=25C, Vfb = 4V Ta=25C, Vfb = 0V 0.70 5.5 3.5 0.90 6.0 5.0 1.1 6.5 6.5 mA V A FOSC FMOD FOSC FMOD F/T Dmax FSDH321 FSDL321 -25C Ta +85C FSDH321 FSDL321 90 2.5 45 1.0 62 71 100 3 50 1.5 5 67 77 110 3.5 55 2.0 10 72 83 kHz kHz % % % VSTART VSTOP VFB=GND VFB=GND 11 7 12 8 13 9 V V Symbol Condition Min. Typ. Max. Unit
Note: 1. These parameters, although guaranteed, are not 100% tested in production 2. These parameters, although guaranteed, are tested in EDS (wafer test) process 3. di/dt = 250mA/uS
6
FSDH321, FSDL321
Comparison Between FSDM311 and FSDx321
Function Soft-Start FSDM311 15mS FSDx321 15mS FSDx321 Advantages * Gradually increasing current limit during soft-start further reduces peak current and voltage component stresses * Eliminates external components used for soft-start in most applications * Reduces or eliminates output overshoot * Smaller transformer * Allows power limiting (constant overload power) * Allows use of larger device for lower losses and higher efficiency. * Reduced conducted EMI * Improve light load efficiency * Reduces no-load consumption * Transformer audible noise reduction * Greater immunity to arcing as a result of build-up of dust, debris and other contaminants
External Current Limit
not applicable
Programmable of default current limit
Frequency Modulation Burst Mode Operation
not applicable Yes-built into controller 7.62mm
1.5KHz @50KHz 3.0KHz @100KHz Yes-built into controller 7.62mm
Drain Creepage at Package
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FSDH321, FSDL321
Typical Performance Characteristics (Control Part)
(These characteristic graphs are normalized at Ta = 25C)
1.20 1.00 Normalized 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
Normalized
1.20 1.00 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
Operating Frequency (Fosc)
Frequency Modulation (FMOD)
1.20 1.00 Normalized 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
1.20 1.00 Normalized 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
Maximum duty cycle (Dmax)
Operating supply current (Iop)
1.20 1.00
Normalized
1.20 1.00 0.80 0.60 0.40 0.20 0.00
Nomalized
0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
-50
0
50 T emp[ ]
100
150
Start Threshold Voltage (Vstart)
Stop Threshold Voltage (Vstop)
8
FSDH321, FSDL321
Typical Performance Characteristics (Continued)
1.20 1.00 Normalized
Normalized
1.20 1.00 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
Feedback Source Current (Ifb)
Peak current limit (ILIM)
1.20 1.00 Normalized 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150 Normalized
1.20 1.00 0.80 0.60 0.40 0.20 0.00 -50 0 50 T emp[ ] 100 150
Start up Current (Istart)
Startup Charging Current (Ich)
1.20 1.00 Normalized
1.20 1.00 Normalized 0.80 0.60 0.40 0.20 0.00
-50 0 50 T emp[ ] 100 150
0.80 0.60 0.40 0.20 0.00
-50
0
50 Temp[]
100
150
Burst peak current (Iburst)
Over Voltage Protection (Vovp)
9
FSDH321, FSDL321
Functional Description
1. Startup : In previous generations of Fairchild Power Switches (FPSTM) the Vstr pin had an external resistor to the DC input voltage line. In this generation the startup resistor is replaced by an internal high voltage current source and a switch that shuts off when 15mS goes by after the supply voltage, Vcc, gets above 12V. The source turns back on if Vcc drops below 8V.
Vcc 2uA Vref 0.9mA
Vo
Vfb
FB 3
Cfb
OSC
D1
D2 28R Vfb* R Gate driver
431
Vin,dc Istr
VSD
OLP
Figure 5. Pulse width modulation (PWM) circuit
Vstr Vcc
UVLO <8V on 15m S After UVLO start(>12V) off
J-FET
Figure 4. High voltage current source
2. Feedback Control : The FSDx321 employs current mode control, shown in figure 5. An opto-coupler (such as the H11A817A) and shunt regulator (such as the KA431) are typically used to implement the feedback network. Comparing the feedback voltage with the voltage across the Rsense resistor plus an offset voltage makes it possible to control the switching duty cycle. When the reference pin voltage of the KA431 exceeds the internal reference voltage of 2.5V, the H11A817A LED current increases, thus pulling down the feedback voltage and reducing the duty cycle. This event typically happens when the input voltage is increased or the output load is decreased.
4. Protection Circuit : The FPSTM has several protective functions such as over load protection (OLP), over voltage protection (OVP), abnormal over current protection (AOCP), under voltage lock out (UVLO) and thermal shutdown (TSD). Because these protection circuits are fully integrated inside the IC without external components, the reliability is improved without increasing cost. Once the fault condition occurs, switching is terminated and the Sense FET remains off. This causes Vcc to fall. When Vcc reaches the UVLO stop voltage, 8V, the protection is reset and the internal high voltage current source charges the Vcc capacitor via the Vstr pin. When Vcc reaches the UVLO start voltage,12V, the FPSTM resumes its normal operation. In this manner, the auto-restart can alternately enable and disable the switching of the power Sense FET until the fault condition is eliminated.
3. Leading edge blanking (LEB) : At the instant the internal Sense FET is turned on, there usually exists a high current spike through the Sense FET, caused by the primary side capacitance and secondary side rectifier diode reverse recovery. Excessive voltage across the Rsense resistor would lead to incorrect feedback operation in the current mode PWM control. To counter this effect, the FPSTM employs a leading edge blanking (LEB) circuit. This circuit inhibits the PWM comparator for a short time (TLEB) after the Sense FET is turned on.
4.1 Over Load Protection (OLP) : Overload is defined as the load current exceeding a pre-set level due to an unexpected event. In this situation, the protection circuit should be activated in order to protect the SMPS. However, even when the SMPS is in the normal operation, the over load protection circuit can be activated during the load transition. In order to avoid this undesired operation, the over load protection circuit is designed to be activated after a specified time to determine whether it is a transient situation or an overload situation. In conjunction with the Ipk current limit pin (if used) the current mode feedback path would limit the current in the Sense FET when the maximum PWM duty cycle is attained. If the output consumes more than this maximum power, the output voltage (Vo) decreases below the set voltage. This reduces the current through the opto-coupler LED, which also reduces the opto-coupler transistor current, thus
increasing the feedback voltage (Vfb). If Vfb exceeds 3V, the feedback input diode is blocked and the 5uA Idelay current source starts to charge Cfb slowly up to Vcc. In this condition, Vfb continues increasing until it reaches 6V, when the switching operation is terminated as shown in figure 6. The delay time for shutdown is the time required to charge Cfb from 3V to 6V with 5uA.
10
FSDH321, FSDL321
Vcc
8V OLP 6V
enabled and monitors the current through the sensing resistor. The voltage across the resistor is then compared with a preset AOCP level. If the sensing resistor voltage is greater than the AOCP level, pulse by pulse AOCP is triggered regardless of uncontrollable LEB time. Here, pulse by pulse AOCP stops Sense FET within 350nS after it is activated.
FPS switching
Following Vcc
3V Delay current (5uA) charges the Cfb
t1
t1 = - 1 RC
t2
In (1 -
fb
t3
t4
t
V ( t 1) ); V ( t1) = 3V , R = 2 . 8 K , C fb = C R
fb _ fig . 2
t 2 = C fb
(V (t1 + t 2) - V (t1)) ; I delay = 5uA,V (t1 + t 2) - V (t1) = 3V I delay
Figure 6. Over load protection
4.2 Thermal Shutdown (TSD) : The Sense FET and the control IC are integrated, making it easier for the control IC to detect the temperature of the Sense FET. When the temperature exceeds approximately 140C, thermal shutdown is activated.
4.4 Over Voltage Protection (OVP) : In case of malfunction in the secondary side feedback circuit, or feedback loop open caused by a defect of solder, the current through the opto-coupler transistor becomes almost zero. Then, Vfb climbs up in a similar manner to the over load situation, forcing the preset maximum current to be supplied to the SMPS until the over load protection is activated. Because excess energy is provided to the output, the output voltage may exceed the rated voltage before the over load protection is activated, resulting in the breakdown of the devices in the secondary side. In order to prevent this situation, an over voltage protection (OVP) circuit is employed. In general, Vcc is proportional to the output voltage and the FPSTM uses Vcc instead of directly monitoring the output voltage. If VCC exceeds 19V, OVP circuit is activated resulting in termination of the switching operation. In order to avoid undesired activation of OVP during normal operation, Vcc should be properly designed to be below 19V.
4.3 Abnormal Over Current Protection (AOCP) :
PW M COM PARATOR
Vfb LEB
CLK Out Driver Drain
Vsense
AOCP COMPARATOR
S R
Q
5. Soft Start : The FPSTM has an internal soft start circuit that increases the feedback voltage together with the Sense FET current slowly after it starts up. The typical soft start time is 15msec, as shown in figure 8, where progressive increments of Sense FET current are allowed during the start-up phase. The pulse width to the power switching device is progressively increased to establish the correct working conditions for transformers, inductors, and capacitors. The voltage on the output capacitors is progressively increased with the intention of smoothly establishing the required output voltage. It also helps to prevent transformer saturation and reduce the stress on the secondary diode.
VAOCP
Rsense
Drain current [A]
Figure 7. AOCP Function & Block
0.7A
Even though the FPSTM has OLP (Over Load Protection) and current mode PWM feedback, these are not enough to protect the FPSTM when a secondary side diode short or a transformer pin short occurs. In addition to start-up, softstart is also activated at each restart attempt during autorestart and when restarting after latch mode is activated. The FPSTM has an internal AOCP (Abnormal Over Current Protection) circuit as shown in figure 7. When the gate turn-on signal is applied to the power Sense FET, the AOCP block is
0.4A
Tss
11
FSDH321, FSDL321
5V
D R A IN
Burst Operation Feedback
Burst Operation
Normal Operation
S W IT C H OFF GND
0.5V 0.5V
I_ o v e r
Rsense
0.3V 0.35V
Current waveform Switching OFF Switching OFF
Figure 8. Soft Start Function Figure 10. Circuit for Burst Operation
6. Burst operation :In order to minimize power dissipation in standby mode, the FPSTM enters burst mode operation. 7. Frequency Modulation : EMI reduction can be accomplished by modulating the switching frequency of a switched power supply. Frequency modulation can reduce EMI by spreading the energy over a wider frequency range than the band width measured by the EMI test equipment. The amount of EMI reduction is directly related to the depth of the reference frequency. As can be seen in Figure 11, the frequency changes from 97KHz to 100KHz (from 48.5KHz to 51.5KHz ; FSDL321)in 4mS for the FSDH321. Frequency modulation allows the use of a cost effective inductor instead of an AC input mode choke to satisfy the requirements of world wide EMI limits.
+
0.35V/0.5V 0.3/0.5V 0.5V 0.5V
Vcc IB_PEAK
Vcc Idelay
Vcc IFB
FB 3
Normal
PWM
Burst
2.5R R
MOSFET Current
Figure 9. Circuit for Burst operation
Internal Oscillator
103kHz
As the load decreases, the feedback voltage decreases. As shown in figure 10, the device automatically enters burst mode when the feedback voltage drops below VBURH(500mV). Switching still continues but the current limit is set to a fixed limit internally to minimize flux density in the transformer. The fixed current limit is larger than that defined by Vfb = VBURH and therefore, Vfb is driven down further. Switching continues until the feedback voltage drops below VBURL(350mV). At this point switching
Drain to Source voltage
Drain to
stops and the output voltages start to drop at a rate dependent on the standby current load. This causes the feedback voltage to rise. Once it passes VBURH(500mV) switching resumes. The feedback voltage then falls and the process repeats. Burst mode operation alternately enables and disables switching of the power Sense FET thereby reducing switching loss in
Standby mode.
Vds Waveform 97kHz 100kHz 103kHz Turn-on Turn-off 6kHz
Figure 11. Frequency Modulation Waveform for FSDH321
12
FSDH321, FSDL321
8. Adjusting Current limit function: As shown in fig 12, a combined 2.8K internal resistance is connected into the non-inverting lead on the PWM comparator. A external resistance of Y on the current limit pin forms a parallel resistance with the 2.8K when the internal diodes are biased by the main current source of 900uA.
5uA Feed Back
900uA
3 2 K
Current Limit PWM comparator
4 AK
0.8 K
Rsense SenseFET Sense
Figure 12. Peak current adjustment
For example, FSDH321 has a typical Sense FET current limit (ILIM) of 0.7A. The Sense FET current can be limited to 0.5 by inserting a k between the current limit pin and ground which is derived from the following equations: 0.7: 0.5 = 2.8K : XK , X = 2K, Since X represents the resistance of the parallel network, Y can be calculated using the following equation: Y = X / (1 - (X/2.8K)) ; Y = 7K
13
FSDH321, FSDL321
Typical application circuit
1. PC Auxiliary Power Circuit (10W Output Power)
140~375 VDC INPUT R102 100k 1W C101 10nF 630V
T1 EE1625 1 10
D201 SB360 C201 1000uF 16V 7
L201 10uH
C203 470uF 16V
5V (+/-5%) 2A
2
R101 680k 1W
D101 UF 4007 3 M Vcc D102 1N4937 C102 47uF 50V R104 10 5 R103 10
5 IC101 FSDx321 3 Vstr Drain 6,7,8 Vfb Vcc C104 22nF GND 1 2
4
D103 1N4937
C103 10uF 50V
6 PC301 H11A817A
R202 330 R203 2k C202 100nF
R201 1k
C301 2.2nF
IC201 KA431
R204 2k
10W PC Auxiliary Power Circuit
10W PC Auxiliary Power, 150~375VDC Input Power supply: It shows a auxiliary power for PC. Efficiency at 10W, 150/ 375VDC is 70%. The PC application has the standard of standby power consumption, under 1W at the output load, 0.5W and height input voltage, 230VAC. For this the FSDH321 also has the burst operating function like the any other green mode FPS like FSDM0265RN or FSDM0365RN and so on. This skill reduces the MOSFET switching numbers and power MOSFET switching loss. This design takes advantage of self protection without external components and high switching frequency, 100kHz. The frequency makes using a small size transformer core possible. The EE16 or EE1625 can be used for 10W application. This is achieved by preventing the green FPS from switching when the input voltage goes below a level needed to maintain output regulation, and keeping it off until the input voltage goes above the under-voltage threshold, when the AC is turned on again. For example with the resistor, R101, 680k, the threshold voltage is around 150VAC(210VDC) at the room temperature. Leakage inductance clamping is provided by R102 and 14
C101, keeping the DRAIN voltage below 650 V under all conditions. And R102 dissipates power to prevent rising of DRAIN Voltage caused by leakage inductance. The frequency modulation feature of FSDH321 allows the circuit shown to meet CISPR2AB with simple EMI filtering. The secondary is rectified and smoothed by D201. Similarly D102 and D103 are also rectifiers for main power control IC and FSDH321 respectively. The 5V output voltage require two capacitors in parallel to meet the ripple current requirement. Switching noise filtering is provided by L201. The output is regulated by the reference (TL431) voltage in secondary. It is sensed via R203 and R204. Resistor R201 provides bias for TL431 and R202 sets the overall DC gain. R2012, C202 and R203 provide loop compensation.
FSDH321, FSDL321
2. Transformer Specification (10W Output Power)
1. Schematic Diagram
EE1625 N p/2 1 10
Na Np/2 NM Vcc
N p/2 2 3 4 5 6
N 5V
9 8
N M Vcc Na
N5V
7
Np/2
2. Winding Specification
P in ( S ! F ) N p /2 3!2 W ire 0 .1 5 x 1 T u rn s 80 W in d in g M e th o d S o le n o id w in d in g
In s u la tio n : P o ly e s te r T a p e t = 0 .0 5 0 m m , 3 L a y e rs N 5V 10 ! 7 0 .5 5 x 1 12 S o le n o id w in d in g
In s u la tio n : P o ly e s te r T a p e t = 0 .0 5 0 m m , 3 L a y e rs N M VCC 4!6 0 .2 0 x 1 40 S o le n o id w in d in g
In s u la tio n : P o ly e s te r T a p e t = 0 .0 5 0 m m , 3 L a y e rs N p /2 2!1 0 .1 5 x 1 80 S o le n o id w in d in g
In s u la tio n : P o ly e s te r T a p e t = 0 .0 5 0 m m , 3 L a y e rs Na 5!6 0 .2 0 x 1 34 S o le n o id w in d in g
O u te r In s u la tio n : P o ly e s te r T a p e t = 0 .0 5 0 m m , 3 L a y e rs
3. Electric Specification and Core and Bobbin
P in In d u c ta n c e L e a ka g e C o re B o b b in 1-3 1-3
Spec. 1 .8 m H 100uH EE1625 EE1625
R e m a rk 1 kH z, 1 V 2 n d s id e a ll s h o rt
15
FSDH321, FSDL321
Layout Considerations
SURFACE MOUNTED COPPER AREA FOR HEAT SINKING
DC_link Capacitor
#1 : GND #2 : VCC #3 : Vfb #4 : Ipk #5 : Vstr #6 : Drain #7 : Drain #8 : Drain Y1CAPACITOR
-+ DC OUT
Figure 13. Layout Considerations for FSDx321 using 8DIP
16
FSDH321, FSDL321
Package Dimensions
8DIP
17
FSDH321, FSDL321
Package Dimensions (Continued)
8LSOP
18
FSDH321, FSDL321
Ordering Information
Product Number FSDH321 FSDL321 FSDH321L FSDL321L Package 8DIP 8DIP 8LSOP 8LSOP Marking Code DH321 DL321 DH321 DL321 BVDSS 650V 650V 650V 650V FOSC 100KHz 50KHz 100KHz 50KHz RDS(on) 14 14 14 14
19
FSDH321, FSDL321
DISCLAIMER FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS. LIFE SUPPORT POLICY FAIRCHILD'S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and (c) whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury of the user.
www.fairchildsemi.com 10/1/04 0.0m 001 Stock#DSxxxxxxxx 2004 Fairchild Semiconductor Corporation
2. A critical component in any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness.

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